Ultrawide-band communication system and method

ABSTRACT

An impulse radio communications system using one or more subcarriers to communicate information from an impulse radio transmitter to an impulse radio receiver. The impulse radio communication system is an ultrawide-band time domain system. The use of subcarriers provides impulse radio transmissions added channelization, smoothing and fidelity. Subcarriers of different frequencies or waveforms can be used to add channelization of impulse radio signals. Thus, an impulse radio link can communicate many independent channels simultaneously by employing different subcarriers for each channel. The impulse radio uses modulated subcarrier(s) for time positioning a periodic timing signal or a coded timing signal. Alternatively, the coded timing signal can be summed or mixed with the modulated subcarrier(s) and the resultant signal is used to time modulate the periodic timing signal. Direct digital modulation of data is another form of subcarrier modulation for impulse radio signals. Direct digital modulation can be used alone to time modulate the periodic timing signal or the direct digitally modulated the periodic timing signal can be further modulated with one or more modulated subcarrier signals. Linearization of a time modulator permits the impulse radio transmitter and receiver to generate time delays having the necessary accuracy for impulse radio communications.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.10/196,121, filed Jul. 12, 2002, now U.S. Pat. No. 6,847,675, which is acontinuation of U.S. patent application Ser. No. 09/037,704, filed Mar.10, 1998, now U.S. Pat. No. 6,430,208, which is a continuation of U.S.patent application Ser. No. 08/949,144, filed Oct. 10, 1997, now U.S.Pat. No. 5,995,534, which is a division of U.S. patent application Ser.No. 08/309,973, filed Sep. 20, 1994, now U.S. Pat. No. 5,677,927, thespecifications of which are incorporated herein by reference in theirentireties.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to the field of communications, and moreparticularly, the present invention relates to ultrawide-band impulsecommunication systems and methods employing subcarriers.

2. Background Art

Designers of radio technology for personal communications devices,medical and military devices, and the like, are currently faced withseveral development challenges. Low power consumption, reuse ofavailable spectrum, channelization and cost are four of the main issues.

These issues are addressed in part by an emerging, revolutionarytechnology called impulse radio communications (hereafter called impulseradio). Impulse radio was first fully described in a series of patents,including U.S. Pat. Nos. 4,641,317 (issued Feb. 3, 1987), 4,813,057(issued Mar. 14, 1989), 4,979,186(issued Dec. 18, 1990) and 5,363,108(issued Nov. 8, 1994), all to Larry W. Fullerton. These patent documentsare incorporated herein by reference.

Basic impulse radio transmitters emit short Gaussian monocycle pulseswith tightly controlled average pulse-to-pulse interval. Impulse radiosystems use pulse position modulation. Pulse position modulation is aform of time modulation in which the value of each instantaneous sampleof a modulating signal is caused to modulate the position in time of apulse.

For impulse radio communications, the pulse-to-pulse interval is variedon a pulse-by-pulse basis by two components: an information componentand a pseudo-random code component. Spread spectrum systems make use ofpseudo-random codes to spread the normally narrowband information signalover are latively wide band of frequencies. A spread spectrum receivercorrelates these signals to retrieve the original information signal.Unlike spread spectrum systems, the pseudo-random code for impulse radiocommunications is not necessary for energy spreading because themonocycle pulses themselves have an inherently wide informationbandwidth (information bandwidth, hereafter called bandwidth, is therange of frequencies within which performance, with respect to somecharacteristics, falls within specific limits). Instead, thepseudo-random code is used for channelization, energy smoothing in thefrequency domain, and jamming resistance.

The impulse radio receiver is a homodyne receiver with a crosscorrelator front end. The front end coherently converts anelectromagnetic pulse train of monocycle pulses to a baseband signal ina single stage. (The baseband signal is the basic information channelfor the basic impulse radio communications system, and is also referredto as the information bandwidth.) The data rate of the impulse radiotransmission is only a fraction of the periodic timing signal used as atime base. Each data bit time position modulates many pulses of theperiodic timing signal. This yields a modulated, coded timing signalthat comprises a train of identical pulses for each single data bit. Thecross correlator of the impulse radio receiver integrates multiplepulses to recover the transmitted information.

As with all aspects of the electronics field, what is desired are stillsmaller, lower power and more flexible systems. However, generallyaccepted principles in continuous wave (CW) radio technology do notreadily lend themselves to time domain systems, such as impulse radio.

Descriptions of some of the basic concepts discussed below are found ina number of references, including Robert C. Dixon, Spread SpectrumSystems (John Wiley & Sons, Inc., New York, 1984, 2nd ed.); and Don J.Torrieri, Principles of Military Communication Systems (Artech House,Inc., Dedham Mass., 1982, 3rd ed.).

BRIEF SUMMARY OF THE INVENTION

The impulse radio communications system according to the presentinvention uses one or more subcarriers to communicate information froman impulse radio transmitter to an impulse radio receiver. Three impulseradio communications system embodiments are described, including: a onechannel system, a two channel system and a three or more channel system.Typical radio frequency impulse radio communications system applicationsinclude cellular telephones, wireless telephones, wireless PBXs/Localarea networks, and the like.

The impulse radio communication system is an ultrawide-band time domainsystem. Operation in the time domain is in accordance with generalimpulse radio theories discussed below in section II. The use ofsubcarriers provides impulse radio transmissions added channelization,smoothing and fidelity. Subcarriers of different frequencies orwaveforms can be used (simultaneously) to add channelization of impulseradio signals. Thus, an impulse radio link can communicate manyindependent channels simultaneously by employing different subcarriersfor each channel.

There are three impulse radio transmitter embodiments. The first andsecond transmitter embodiments comprise a subcarrier generator andmodulator that uses one or more information signals to modulate aperiodic timing signal.

According to the first embodiment, coding of the impulse radio signalsis achieved by coding the periodic timing signal before it is timemodulated by the modulated subcarrier signal.

According to the second embodiment, coding of the impulse radio signalsis achieved by coding a modulated subcarrier signal before it is used totime modulate the periodic timing signal.

The third transmitter embodiment comprises a subcarrier generator andmodulator that uses one or more information signals to modulate aperiodic timing signal in combination with direct digital modulation ofa digital data signal. In this embodiment, the modulated subcarriersignal is used to time modulate the direct digitally modulated signal.

The impulse radio transmitter generally comprises a time base thatgenerates a periodic timing signal. The time base comprises a voltagecontrolled oscillator, or the like, having sub-nanosecond timingrequirements. The periodic timing signal is supplied to a code sourceand to a code time modulator. The code source comprises a storage devicefor storing nearly orthogonal pseudo-random noise (PN) codes and meansfor outputting the PN codes as a code signal. The code source monitorsthe periodic timing signal to permit the code signal to be synchronizedto the code time modulator. In one embodiment, the code time modulatoruses the code signal to modulate the periodic timing signal forchannelization and smoothing of a final emitted impulse radio signal.The output of the code time modulator is called the coded timing signal.

The coded timing signal is supplied to a subcarrier time modulator forinformation modulation thereof. Prior impulse systems usednon-subcarrier, baseband modulation. In other words, the informationitself was used for modulation. In the present invention, however, aninformation source supplies an information signal to a subcarriergenerator and modulator. The information signal can be any type ofintelligence, including digital bits representing voice, data, imagery,or the like, analog signals, or complex signals.

The subcarrier generator and modulator of the present inventiongenerates a modulated subcarrier signal that is modulated by theinformation signal, and supplies the modulated subcarrier signal to thesubcarrier time modulator. Thus, the modulated subcarrier signal is usedby the subcarrier time modulator to modulate the carrier, which in thiscase is the coded timing signal. Modulation of the coded timing signalby the subcarrier time modulator generates a modulated, coded timingsignal that is sent to an output stage.

The output stage uses the modulated, coded timing signal as a trigger togenerate monocycle pulses. In a radio frequency embodiment, themonocycle pulses are sent to a transmit antenna via a transmission linecoupled thereto. The monocycle pulses are converted into propagatingelectromagnetic pulses by the transmit antenna. The emitted signalpropagates to an impulse radio receiver through a propagation medium,such as air in a radio frequency embodiment. In the preferredembodiment, the emitted signals are wide-band or ultrawide-band signals.The spectrum of the emitted signals can be modified by filtering of themonocycle pulses. This filtering will cause each monocycle pulse to havemore zero crossings in the time domain. In this case, the impulse radioreceiver must use a similar waveform in the cross correlator to beefficient.

There are several impulse radio receiver embodiments. Each impulse radioreceiver generally comprises a cross correlator, a decode source, adecode timing modulator and adjustable time base and a subcarrierdemodulator.

The decode source generates a decode control signal corresponding to thePN code used by an impulse radio transmitter communicating an impulseradio signal. The adjustable time base generates a periodic timingsignal that comprises a train of template signal pulses having waveformssubstantially equivalent to each pulse of the received (impulse radio)signal.

The decode timing modulator uses the decode control signal to positionin time a periodic timing signal to produce a decode signal. The decodesignal is thus matched in time to the known PN code of the transmitterso that the received signal can be detected in the cross correlator.

The decode signal is used to produce a template signal having a waveformdesigned to match the received signal. The template signal is positionedin time according to the known PN code of the transmitter and is thencross correlated with the received signal. Successive cross correlationoutput signals are integrated to recover the impulse radio signal out ofthe noise. Once retrieved in this manner, the signal is demodulated toremove the subcarrier and yield the information signal.

The baseband signal is also input to a lowpass filter. A control loopcomprising the lowpass filter is used to generate an error signal toprovide minor phase adjustments to the adjustable time base to timeposition the periodic timing signal in relation to the position of thereceived signal.

In a preferred embodiment, a subcarrier in an impulse radio translates(or shifts) the baseband signals to a higher frequency. The subcarriergeneration and modulator generates a signal that is modulated by theinformation signal by frequency modulation (FM) techniques, amplitudemodulation (AM), phase modulation, frequency shift keying (FSK), phaseshift keying (PSK), pulsed FM, or the like.

Other non-sinusoidal and/or non-continuous waveforms can also beemployed as subcarriers in connection with the present invention. Themodulated subcarrier signal is used to time shift the position of thepulses of the coded timing signal or the periodic timing signal. Thus,the signal that triggers the output stage is a train of pulse positionmodulated pulses. In another embodiment, direct digital modulation usingManchester encoding is employed as a subcarrier. Combination of thesesubcarrier techniques is also described.

The effect of using the cross correlation function for the modulationtransfer function is to cause the output of the receiver to be anon-linear function of the amplitude of the input. For basebandmodulation, this is undesirable. However, for subcarriers, such as FM,AM, FSK, PSK and Manchester, the harmonics are filtered therebyeliminating any distortion. Such filtering can not remove harmonics whenbaseband modulation is used, because the harmonics stay at baseband, andthus the signal is irrecoverable.

The addition of subcarriers also provides added fidelity in the form ofmore bandwidth and better signal-to-noise, compared to basebandmodulation alone. This benefit is attributed to the fact that thesubcarrier inherently renders the information more impervious to noise.The subcarrier embodiments provide less signal compression, and lowersignal distortion by reducing baseband noise for high reliability voice,data and/or imagery communications.

The linearity requirements for the modulation using the cross correlatorare greatly relaxed by using the subcarrier technique of the presentinvention. The use of a subcarrier for impulse radios also improvesharmonic distortion due to a non-linear modulation transfer function,compared to baseband modulation. Modulation transfer characteristicshave to be extremely linear in order to successfully transfer lowdistortion speech or music. This is very difficult to achieve in anon-subcarrier baseband impulse system.

The foregoing and other features and advantages of the present inventionwill be apparent from the following more particular description of thepreferred embodiments of the invention, as illustrated in theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES

FIGS. 1A and 1B show a 2 GHz center frequency monocycle pulse in thetime and frequency domains, respectively, in accordance with the presentinvention.

FIGS. 2A and 2B are illustrations of a 1 mpps system with 1 ns pulses inthe time and frequency domains, respectively, in accordance with thepresent invention.

FIG. 3 illustrates a modulating signal that changes the pulse repetitioninterval (PRI) in proportion to the modulation in accordance with thepresent invention.

FIG. 4 is a plot illustrating the impact of pseudo-random dither onenergy distribution in the frequency domain in accordance with thepresent invention.

FIG. 5 illustrates the result of a narrowband sinusoidal (interference)signal overlaying an impulse radio signal in accordance with the presentinvention.

FIG. 6 shows the “cross correlator transfer function of an impulse radioreceiver in accordance with the present invention.

FIG. 7 illustrates impulse radio multipath effects in accordance withthe present invention.

FIG. 8 illustrates the phase of the multipath pulse in accordance withthe present invention.

FIG. 9 shows a representative block diagram of an impulse radioelectrical system using one subcarrier channel in accordance with thepresent invention.

FIG. 10 shows an impulse radio transmitter of an impulse radiocommunication system in accordance with the present invention.

FIG. 11 shows another embodiment of the impulse radio transmitter inaccordance with the present invention.

FIG. 12 shows another transmitter embodiment in accordance with thepresent invention.

FIG. 13 shows a still further alternate embodiment in accordance withthe present invention.

FIG. 14 shows an impulse radio receiver in accordance with the presentinvention.

FIG. 15 shows a representative plot of a pulse corresponding to thereceived signal in connection with the receiver 1400 in accordance withthe present invention.

FIG. 16 illustrates the cross correlation process in accordance with thepresent invention.

FIG. 17 shows a representative illustration of an impulse radiotransmitter having three subcarrier generator/modulators in accordancewith the present invention.

FIG. 18 is a representative analog embodiment showing the crosscorrelator followed by plural analog FM demodulation branches inaccordance with the present invention.

FIG. 19 shows a digital embodiment in accordance with the presentinvention.

FIG. 20 is a plot showing delay time (in picoseconds) versus a binary(i.e., numeric) input value for a conventional binary-to-time delaygenerator in accordance with the present invention.

FIG. 21 is a high-level block diagram showing the above linearizationscheme in accordance with the present invention.

FIG. 22 is a functional diagram illustrating linearization ROM 2110 inaccordance with the present invention.

FIG. 23 shows a combined PN code and linearization E-PROM in accordancewith the present invention.

FIG. 24 illustrates a further embodiment of the impulse radio receiverin accordance with the present invention.

FIGS. 25A-25H illustrate time(t) versus voltage plots of various signalsnumbered in FIG. 24 in accordance with the present invention.

FIGS. 25I-25L illustrate frequency versus amplitude plots correspondingto FIGS. 25E-25H in accordance with the present invention.

FIGS. 26 and 27 show exemplary waveforms for pseudo Manchester encodingand decoding, respectively, in accordance with the present invention.

FIG. 28 is a high-level block diagram of operations performed by theimpulse radio receiver to acquire lock in accordance with the presentinvention.

FIG. 29 shows the signal measured at 3 meters as well as ambient signalsin accordance with the present invention.

FIG. 30 shows a curve that illustrates a specific example of theprojected trade-off between free space range and bit rate in accordancewith the present invention.

FIG. 31 shows that it is easy to resolve multipath impulse signals inthe time domain in accordance with the present invention.

DETAILED DESCRIPTION OF THE INVENTION I. Overview

The impulse radio communication system is an ultrawide-band time domainsystem that operates in the time domain and uses one or more subcarriersto provide channelization, smoothing and fidelity. A single impulseradio transmission (e.g., a link) can therefore communicate manyindependent channels simultaneously by employing different subcarriersfor each channel.

The impulse radio transmitter according to the present invention usesmodulated subcarrier(s) for time positioning a periodic timing signal ora coded timing signal. Alternatively, the coded timing signal can bemixed (or summed) with the modulated subcarrier(s) and the resultantsignal used to time modulate the periodic timing signal. Direct digitalmodulation of data is another form of subcarrier modulation for impulseradio signals. Direct digital modulation can be used alone to timemodulate the periodic timing signal or the direct digitally modulatedperiodic timing signal can be further modulated with one or moremodulated subcarrier signals.

Impulse radio technology according to the present invention is widelyapplicable for wireless communications applications. Because impulseradio is not a continuous wave (CW) carrier-based system, the use of asubcarrier is an elegant, counter intuitive addition to the time domainimpulse radio design. Signal-to-noise is improved considerably comparedto non-subcarrier impulse radio transmissions.

At first blush, the addition of a subcarrier to an impulse radiocommunication system would appear superfluous. However, the layering ofsubcarrier modulation over information modulation and PN code smoothingin an impulse radio system yields an elegant result.

Impulse radios generally have: short duration pulses; center frequenciestypically between 50 MHz and 10 gigahertz (GHz); ultrawide bandwidths of100+% of the center frequency; multi-mile ranges with sub-milliwattaverage power levels, even with low gain antennas; extremely low powerspectral densities; lower cost than other sophisticated radio designs,especially spread spectrum systems; and excellent immunity to jammingfrom other systems and to multipath fading.

Additionally, impulse radios have exceptional multipath immunity, theyare relatively simple and less costly to build, especially in comparisonto spread spectrum radios. Impulse radio systems consume substantiallyless power than existing conventional radios. Additionally, impulseradio systems occupy less space than existing portabletelecommunications transceivers.

Because of these characteristics impulse radio is an optimal technologyfor a wide variety of applications, including personal communicationssystems and in-building communications systems.

The following sections II through VIII are a detailed description of thepresent invention.

Section II is directed to technology basics and provides the reader withan introduction to impulse radio concepts, as well as other relevantaspects of communications theory. This section includes subsectionsrelating to Gaussian monocycle pulses, pulse trains of gaussianmonocycle pulses, modulation, coding, and qualitative and quantitativecharacteristics of these concepts.

Section III is directed to the use of subcarriers for impulse radiocommunication systems. This section includes subsections relating to thetheory of operation of subcarriers for the impulse radio transmitter andreceiver. the description is sectioned to describe a one channelembodiment with improvement over baseband alone and a two or moresubcarrier channel embodiment

Section IV is directed to the time modulator that is used for code timedelaying, subcarrier time delaying and a combination of both. Theoperation and structure of several embodiments for using the timemodulator for subcarrier impulse radio communications are described.

Section V is directed to linearization of the time modulator for boththe impulse radio transmitter and receiver. Linearization of the timemodulator permits the impulse radio transmitter and receiver to generatetime delays having the necessary accuracy for impulse radiocommunications.

Section VI is directed to pseudo Manchester coding for modulation ofdigital data using impulse radio communications.

Section VII is directed to a lock acquisition scheme for the impulseradio receiver to acquire and maintain lock of impulse radio signals.

Section VIII describes the performance of impulse radio communicationssystems in the real world with reference to data collected by theinventors based on prototype testing.

II. Technology Basics

This section is directed to technology basics and provides the readerwith an introduction to impulse radio concepts, as well as otherrelevant aspects of communications theory. This sections includessubsections relating to Gaussian monocycle pulses, pulse trains ofgaussian monocycle pulses, modulation, coding, and qualitative andquantitative characteristics of these concepts.

Impulse radio transmitters emit short Gaussian monocycle pulses with atightly controlled average pulse-to-pulse interval. Impulse radiotransmitters use pulse widths of between 20 and 0.1 nanoseconds (ns) andpulse-to-pulse intervals of between 2 and 5000 ns. These narrowmonocycle pulses have inherently wide-band frequency characteristics.

Impulse radio systems uses pulse position modulation, with the actualpulse-to-pulse interval being varied on a pulse-by-pulse basis by twocomponents: an information component and a pseudo-random code component.Unlike spread spectrum systems, the pseudo-random code is not necessaryfor energy spreading (because the impulses themselves are inherentlywide-band), but rather for channelization, energy smoothing in thefrequency domain, and jamming resistance.

The impulse radio receiver is a homodyne receiver with a crosscorrelator front end. The front end coherently converts theelectromagnetic pulse train to a baseband signal in one stage. Theimpulse radio receiver integrates multiple pulses to recover each bit ofthe transmitted information.

II.1 Gaussian Monocycle

The most basic element of impulse radio technology is the practicalimplementation of a Gaussian monocycle, which are also referred toherein as Gaussian monocycle pulses. A Gaussian monocycle is the firstderivative of the Gaussian function. FIGS. 1A and 1B show a 2 GHz centerfrequency (i.e., a 0.5 ns pulse width) monocycle pulse in the time andfrequency domains (see 102 and 104, respectively). (Actual practiceprevents the transmission of a perfect Gaussian monocycle. In thefrequency domain, this results in a slight reduction in the signal'sbandwidth.) These monocycles, which are sometimes called impulses, arenot gated sine waves.

The Gaussian monocycle waveform is naturally a wide bandwidth signal,with the center frequency and the bandwidth completely dependent uponthe pulse's width. In the time domain, the Gaussian monocycle isdescribed mathematically by:

$\begin{matrix}{{V(t)} = {A\frac{\sqrt{2e}}{\tau}t\;{\mathbb{e}}^{{({- \frac{t}{\tau}})}^{2}}}} & (0001)\end{matrix}$

-   -   Where,    -   A is the peak amplitude of the pulse,    -   t is time, and    -   τ (tau) is a time decay constant.

In the frequency domain, the Gaussian monocycle envelope is:

$\begin{matrix}{{V(\omega)} = {A\;\omega\;\tau^{2}\sqrt{2\;\pi\; e}{\mathbb{e}}^{- \frac{\omega^{2}\tau^{2}}{2}}}} & (0002)\end{matrix}$

The center frequency is then:

$\begin{matrix}{{f\; c} = {\frac{1}{2\;\pi\;\tau}\mspace{14mu} H\; z}} & (0003)\end{matrix}$

Relative to c, the 3 dB down points (power) are:f_(lower)=0.319 c: f_(upper)=1.922 c.  (0004)

Thus, the bandwidth is approximately 160% of the center frequency.Because τ (tau) also defines the pulse width, then the pulse widthspecifies both the center frequency and bandwidth. In practice, thecenter frequency of a monocycle pulse is approximately the reciprocal ofits length, and its bandwidth is approximately equal to 1.6 times thecenter frequency. Thus, for the “0.5 ns” pulse shown in FIGS. 1A and 1B:f_(c)=2.0 GHz; Δf_(c)=3.2 GHz.  (0005)

II.2 A Pulse Train

Impulse radio systems use pulse trains, not single pulses, forcommunications. As described in detail below in section III, the impulseradio transmitter produces and outputs a train of pulses for each bit ofinformation.

Prototypes built by the inventors have pulse repetition frequencies ofbetween 0.7 and 10 megapulse per second (mpps, where each megapulse is10⁶ pulses). FIGS. 2A and 2B are illustrations of a 1 mpps system with(uncoded, unmodulated) 1 ns pulses in the time and frequency domains(see 202 and 204, respectively). In the frequency domain, this highlyregular pulse train produces energy spikes (comb lines 204) at onemegahertz intervals; thus, the already low power is spread among thecomb lines 204. This pulse train carries no information and, because ofthe regularity of the energy spikes, might interfere with conventionalradio systems at short ranges.

Impulse radio systems have very low duty cycles so the average powertime domain is significantly lower than its peak power in the timedomain. In the example in FIGS. 2A and 2B, for example, the impulsetransmitter operates 0.1% of the time (i.e., 1 ns per microsecond (μs)).

Additional processing is needed to modulate the pulse train so that theimpulse radio system can actually communicate information. Theadditional processing also smoothes the energy distribution in thefrequency domain so that impulse radio transmissions (e.g., signals)interfere minimally with conventional radio systems.

II.3 Modulation

Amplitude and frequency/phase modulation are unsuitable for thisparticular form of impulse communications; the only suitable choice ispulse position modulation, which allows the use of a matched filter(i.e., cross correlator) in the receiver. As illustrated in FIG. 3, amodulating signal changes the pulse repetition interval (PRI) inproportion to the modulation.

If the modulating signal were to have three levels, the first levelmight shift the generation of the pulse forward in time from the nominalby ∂ picoseconds (ps); the second level might not shift the pulseposition in time from the nominal at all; and the third level mightdelay the pulse by ∂ ps. This would be a digital modulation scheme.Analog modulation would allow continuous deviations between PRI-∂ andPRI+∂. In the impulse radio system the maximum value of ∂ is t/4, wheret=time of the pulse.

In the frequency domain, pulse position modulation distributes theenergy over more frequencies. For example, in the case of a 1 mppssystem if the modulation dither (d) were 100 ps, the PRI is 1,000,000Hertz (Hz) and the additional frequency components are: 999,800.04 Hz,999,900.01 Hz, 1,000,100.01 Hz, and 1,000,200.04 Hz. (Dither is animpulse radio communications term for moving the position of a pulse intime.) Transmitted energy is now distributed among more spikes (comblines) in the frequency domain. If the total transmitted energy remainsconstant, the energy in each frequency spike decreases as the number ofpossible pulse positions increases, thus, in the frequency domain, theenergy is more smoothly distributed.

II.4 Coding for Energy Smoothing and Channelization

Because the receiver is a cross correlator, the amount of time positionmodulation required for one-hundred percent modulation is calculated bythe inverse of f_(c)/4 (where f_(c) is the center frequency). For amonocycle with a center frequency of 1.3 GHz, for example, thiscorresponds to ±157 (ps) of time position modulation. Thespectrum-smoothing effects at this level of time dither is negligible.

Impulse radio achieves optimal smoothing by applying to each pulse a PNcode dither with a much larger magnitude than the modulation dither.FIG. 4 is a plot illustrating the impact of pseudo-random dither onenergy distribution in the frequency domain. FIG. 4, when compared toFIG. 2B, shows the impact of using a 256 position PN code relative to anuncoded signal.

PN dithering also provides for channelization (channelization is aprocedure employed to divide a communications path into a number ofchannels). In an uncoded system, differentiating between separatetransmitters would be very hard. PN codes create channels, if the codesthemselves are relatively orthogonal (i.e., there is low correlationand/or interference between the codes being used).

II.5 Reception and Demodulation

Clearly, if there were a large number of impulse radio users within aconfined area, there might be mutual interference. Further, while the PNcoding minimizes that interference, as the number of users rises, theprobability of an individual pulse from one user's sequence beingreceived simultaneously with a pulse from another user's sequenceincreases. Fortunately, implementations of an impulse radio according tothe present invention do not depend on receiving every pulse. Theimpulse radio receiver performs a correlating, synchronous receivingfunction (at the RF level) that uses a statistical sampling of manypulses to recover the transmitted information.

Impulse radio receivers typically integrate 200 or more pulses to yieldthe demodulated output. The optimal number of pulses over which thereceiver integrates is dependent on a number of variables, includingpulse rate, bit rate, jamming levels, and range.

II.6 Jam Resistance

Besides channelization and energy smoothing, the PN coding also makesimpulse radio highly resistant to jamming from all radio communicationssystems, including other impulse radio transmitters. This is critical asany other signals within the band occupied by an impulse signal act as ajammer to the impulse radio. Since there are no unallocated 1+ GHz bandsavailable for impulse systems, they must share spectrum with otherconventional and impulse radios without being adversely affected. The PNcode helps impulse systems discriminate between the intended impulsetransmission and transmissions from others.

FIG. 5 illustrates the result of a narrowband sinusoidal jamming(interference) signal 502 overlaying an impulse radio signal 504. At theimpulse radio receiver, the input to the cross correlator would includethat narrowband signal 502, as well as the received ultrawide-bandimpulse radio signal 504. Without PN coding, the cross correlator wouldsample the jamming signal 502 with such regularity that the jammingsignals could cause significant interference to the impulse radioreceiver. However, when the transmitted impulse signal is encoded withthe PN code dither (and the impulse radio receiver is synchronized withthat identical PN code dither) it samples the jamming signals randomly.According to the present invention, Integrating over many pulses negatesthe impact of jamming.

In statistical terms, the pseudo-randomization in time of the receiveprocess creates a stream of randomly distributed values with a mean ofzero (for jamming signals). All that is necessary to eliminate theimpact of jammers is to sample over enough pulses (i.e., integrate overa sufficiently large number of pulses) to drive the impact of thejamming signals to zero.

II.7 Processing Gain

Impulse radio is jam resistant because of its large processing gain. Forspread spectrum systems, the definition of processing gain, whichquantifies the decrease in channel interference when wide-bandcommunications are used, is the ratio of the bandwidth of the channel tothe bandwidth of the information signal. For example, a direct sequencespread spectrum system with a 10 kHz information bandwidth and a 16 MHzchannel bandwidth yields a processing gain of 1600 or 32 dB. However,far greater processing gains are achieved with impulse radio systemswhere for the same 10 kHz information bandwidth and a 2 GHz channelbandwidth the processing gain is 200,000 or 53 dB.

The duty cycle (e.g., of 0.5%) yields a process gain of 28.3 dB. (Theprocess gain is generally the ratio of the bandwidth of a receivedsignal to the bandwidth of the received information signal.) Theeffective oversampling from integrating over multiple pulses to recoverthe information (e.g., integrating over 200 pulses) yields a processgain of 28.3 dB. Thus, a 2 GHz divided by a 10 mpps link transmitting 50kilobits per second (kbps) would have a process gain of 49 dB, (i.e.,0.5 ns pulse width divided by a 100 ns pulse repetition interval wouldhave a 0.5% duty cycle, and 10 mpps divided by a 50,000 bps would have200 pulses per bit.)

II.8 Capacity

Theoretical analyses suggests that impulse radio systems can havethousands of voice channels per cell. To understand the capacity of animpulse radio system one must carefully examine the performance of thecross correlator. FIG. 6 shows the “cross correlator transfer function”602. This represents the output value of an impulse radio receiver crosscorrelator for any given received pulse. As illustrated at 604, thecross correlator's output is 0 volts when pulses arrive outside of across correlation window 606. As a received pulse 608 slides through thewindow, the cross correlator output varies. It is at its maximum (e.g.,1 volt) when the pulse is τ/4 ahead of the center of the window (asshown at 610), 0 volts when centered in the window (as shown at 612);and at its minimum (e.g., −1 volt) when it is τ/4 after the center.

When the system is synchronized with the intended transmitter, the crosscorrelator's output has a swing of between ±1 volt (as a function of thetransmitter's modulation). Other in-band transmission would cause avariance to the cross correlator's output value. This variance is arandom variable and can be modelled as a Gaussian white noise signalwith a mean value of 0. As the number of interferers increases thevariance increases linearly. By integrating over a large number ofpulses, the receiver develops an estimate of the transmitted signal'smodulation value. Thus, the:

$\begin{matrix}{{{Variance}\mspace{14mu}{of}\mspace{14mu}{the}\mspace{14mu}{Estimate}} = \frac{N\;\sigma}{\sqrt{Z}}} & (0006)\end{matrix}$

Where

-   -   N=number of interferers,    -   σ is the variance of all the interferers to a single cross        correlation, and    -   Z is the number of pulses over which the receiver integrates to        recover the modulation.

This is a good relationship for a communications system for as thenumber of simultaneous users increases, the link quality degradesgradually (rather than suddenly).

II.9 Multipath and Propagation

Multipath fading, the bane of sinusoidal systems, is much less of aproblem (i.e., orders of magnitude less) for impulse systems than forconventional radio systems. In fact, Rayleigh fading, so noticeable incellular communications, is a continuous wave phenomenon, not an impulsecommunications phenomenon.

In an impulse radio system in order for there to be multipath effects,special conditions must persist. The path length traveled by thescattered pulse must be less than the pulse's width times the speed oflight, and/or successively emitted pulses at the transmitter (in thesequence) arrive at the receiver at the same time.

For the former with a one nanosecond pulse, that equals 0.3 meters orabout 1 foot (i.e., 1 ns×300,000,000 meters/second). (See FIG. 7, in thecase where the pulse traveling “Path 1” arrives one half a pulse widthafter the direct path pulse.)

For the latter with a 1 megapulse per second system that would be equalto traveling an extra 300, 600, 900, etc. meters. However, because eachindividual pulse is subject to the pseudo-random dither, these pulsesare decorrelated.

Pulses traveling between these intervals do not cause self-interference(in FIG. 7, this is illustrated by the pulse traveling Path 2). Whilepulses traveling grazing paths, as illustrated in FIG. 7 by thenarrowest ellipsoid, create impulse radio multipath effects.

As illustrated in FIG. 8 at 802, if the multipath pulse travels one halfwidth of a pulse width further, it increases the power level of thereceived signal (the phase of the multipath pulse will be inverted bythe reflecting surface). If the pulse travels less than one half a pulsewidth further it will create destructive interference, as shown at 804.For a 1 ns pulse, for example, destructive interference will occur ifthe multipath pulse travels between 0 and 15 cm (0 and 6 inches).

Tests of impulse radio systems (including impulse radar tests) suggestthat multipath will not present any major problems in actual operation.Additionally, shorter pulse widths are also envisioned, which willfurther reduce the probability of destructive interference (because thereflected path length required for destructive interference will beshortened).

III. The Subcarrier Invention

This section is directed to the use of subcarriers for impulse radiocommunication systems. This section includes subsections relating to thetheory of operation of subcarriers for the impulse radio transmitter andreceiver. the description is sectioned to describe a one channelembodiment with improvement over baseband alone and a two or moresubcarrier channel embodiment.

III.1 Theory of Operation

According to the present invention, impulse radio has been developed toinclude one or more subcarriers for added channelization, smoothing andfidelity. The following ultrawide-band time domain impulse radiocommunication architectures operate according to the general impulseradio theories discussed above in section II. The following threespecific embodiments will be described: a one channel system, a twochannel system and a three or more channel system.

The three impulse radio receiver embodiments set forth below are used byway of example, not limitation, to describe the present invention andenable those skilled in the relevant arts to make and use the invention.These arts include at least the fields of communications, discreteanalog, digital and integrated circuit design and implementation,digital signal processing and PN code theory. The implementation ofvarious elements and blocks will become evident to those skilled in thepertinent art.

III.2 One Channel with Improvement over Baseband Alone

This section describes an impulse radio communications architectureusing one subcarrier channel that has improved performance over basebandalone. The radio frequency (RF) embodiments of the present invention arethe most common. Typical RF impulse radio system applications includecellular telephones, wireless telephones, wireless PBXs/Local areanetworks, and the like.

Propagation, which is defined as the process by which a signal proceedsfrom a transmitter to a receiver, of RF impulse radio signals istypically through air or space from a transmit antenna to a receiveantenna. This is considered wireless RF impulse radio. The preferredantennas for impulse radio are fully described in U.S. Pat. No.5,363,108.

However, the present invention is also suitable for transmission throughcoaxial cable. In this embodiment, the transmit and receive antennas areeliminated.

A representative block diagram of an impulse radio electrical systemusing one subcarrier channel is shown in FIG. 9. A transmitter 901 and areceiver 903 employing a single subcarrier ultrawide-band impulse radiochannel are depicted. The transmitter 901 and the receiver 903 areseparated by a propagation medium 905, such as air, space, or othermedium cable of propagating ultrawide-band signals. Transmitted impulseradio signals 907 propagate through the propagation medium 905 from thetransmitter 901 to the receiver 903.

III.2.a. Transmitter

A preferred embodiment of an impulse radio transmitter 901 of an impulseradio communication system having one subcarrier channel will now bedescribed with reference to FIG. 10.

The transmitter 901 comprises a time base 1002 that generates a periodictiming signal 1004. The time base 1002 comprises a voltage controlledoscillator, or the like, having a high timing accuracy on the order ofpicoseconds. The voltage control to adjust the VCO center frequency isset at calibration the desired center frequency used to define thetransmitter's non-divided pulse repetition rate. The periodic timingsignal 1004 is supplied to a code source 1006 and to a code timemodulator 1008.

The code source 1006 comprises a storage device such as a random accessmemory (RAM), read only memory (ROM), or the like, for storingorthogonal PN codes and for outputting the PN codes as a code signal1010. Alternatively, maximum length shift registers can be used togenerate the PN codes. Code source 1006 monitors the periodic timingsignal 1004 to permit the code signal 1010 to be synchronized to thecode time modulator 1008. The code time modulator 1008 uses the codesignal 1010 to modulate the periodic timing signal 1004 forchannelization and smoothing of a final emitted signal 1012. The outputof the code time modulator 1008 is called coded timing signal 1014.

The coded timing signal 1014 is supplied to a subcarnier time modulator1016 for information modulation thereof. In prior impulse systems, theinformation modulation was done by using the information itself as themodulating source. In the present invention, however, an informationsource 1018 supplies an information signal 1020 to a subcarriergenerator and modulator 1022. The information signal 1020 can be anytype of intelligence, including digital bits representing voice, data,imagery, or the like, analog signals, or complex signals. Both the codedtiming signal 1014 and the subcarrier time modulator 1016 can beimplemented using voltage, current or digital sources as modulationinputs, as would be apparent to a person skilled in the relevant art.

As defined by Dixon, a subcarrier is “a carrier, modulated withinformation separate from carrier modulation, which in turn modulates acarrier.” The subcarrier generator and modulator 1022 of the presentinvention generates a modulated subcarrier signal 1024 which ismodulated by the information signal 1020, and supplies the modulatedsubcarrier signal 1024 to the subcarrier time modulator 1016. Thus, themodulated subcarrier signal 1024 is used by the subcarrier timemodulator 1016 to modulate the carrier, which in this case is the codedtiming signal 1014. Modulation of the coded timing signal 1014 by thesubcarrier time modulator 1016 generates a modulated, coded timingsignal 1026 that is sent to an output stage 1028.

The output stage 1028 uses the modulated, coded timing signal 1026 as atrigger to generate electrical monocycle pulses. The electricalmonocycle pulses are sent to a transmit antenna 1030 via a transmissionline 1032 coupled thereto. The electrical monocycle pulses are convertedinto propagating electromagnetic pulses by the transmit antenna 1030. Inthe present embodiment, the electromagnetic pulses are called theemitted signal 1012, and propagate to an impulse radio receiver (notshown) through a propagation medium 905, such as air in a radiofrequency embodiment. In the preferred embodiment, the emitted signal(s)1012 is wide-band or ultrawide-band signals. However, the emittedsignal(s) 1012 can be spectrally modified by filtering of the monocyclepulses by optional bandpass filter 1029. This bandpass filtering willcause each monocycle pulse to have more zero crossings in the timedomain. In this case, the impulse radio receiver must use a similarwaveform in the cross correlator to be efficient.

The addition of the subcarrier generation and modulation “stage” 1022 tothe impulse radio transmitter 901 has many benefits. The subcarriermodulated by the information signal provides additional channelizationand smoothing to the system permitting the addition of many new,distinct impulse radio channels. The addition of subcarriers alsoprovides added fidelity in the form of more bandwidth and bettersignal-to-noise to the information signal 1020, compared to basebandmodulation alone.

The use of a subcarrier for impulse radios also improves harmonicdistortion due to a non-linear modulation transfer function, compared tobaseband modulation. The non-linear modulation transfer function isdescribed below in connection with the cross correlation processperformed by the impulse radio receiver.

Because impulse radio is not a CW carrier-based system, the use of asubcarrier is an elegant, counter intuitive addition to the time domainimpulse radio design. Signal-to-noise is improved by 5-20 dB (dependingon signal-to-noise of the narrow pulse modulated carrier) compared tonon-subcarrier impulse radio transmissions.

Using a subcarrier in an impulse radio translates (or shifts) thebaseband signals to a higher frequency. In a preferred embodiment, thesubcarrier generation and modulator 1022 generates a signal that ismodulated by the information signal 1020 by frequency modulation (FM)techniques, amplitude modulation (AM), phase modulation, frequency shiftkeying (FSK), phase shift keying (PSK), pulsed FM, or the like. Inanother embodiment, direct digital modulation is employed as asubcarrier technique. In this alternate embodiment, Manchester encodingof digital data produces a digital modulated subcarrier signal 1024. Thesubcarrier time modulator 1016 uses the modulated subcarrier 1024 topulse position modulate the coded timing signal 1014.

Other non-sinusoidal and/or non-continuous waveforms can also beemployed as subcarriers in connection with the present invention. Themodulated subcarrier signal 1024 is used by the subcarrier timemodulator 1016 to time shift the position of the pulses of the codedtiming signal 1014. Thus, the signal that triggers the output stage (inthis case the modulated, coded timing signal 1026) is a train of pulseposition modulated pulses.

Subcarriers of different frequencies or waveforms can be used to addchannelization of impulse radio signals. Thus, an impulse radio link cancommunicate many independent channels simultaneously by employingdifferent subcarriers for each channel.

To illustrate this, consider two separate pairs of impulse radio usersoperating with the same PN codes. A first pair of users communicate withimpulse radios having the subcarrier generator/modulators 1022generating one sine wave subcarrier of a first discrete frequency. Asecond pair of users communicate with separate impulse radios having thesubcarrier generator/modulator 1022 generating one sine wave subcarrierof a second discrete frequency, separate from the first frequency. Eachuser pair can have isolated communications from the other by configuringthe impulse radio receivers of the two pairs (as discussed below) toreproduce only the information conveyed by the appropriate subcarrierfrequency. In view of this illustration, many additional impulse radiochannels are available by using the impulse radio subcarrier technique.

Alternatively, the two pairs of impulse radio users could have isolatedcommunications if each pair used different PN codes and the samesubcarriers. Additionally, channelization can be achieved by having setsof radios operate at different pulse repetition rates, independent of PNcodes and/or subcarrier.

A result of the novel subcarrier stage is enhanced fidelity of theinformation channel. This benefit is attributed to the fact that thesubcarrier inherently renders the information more impervious to noise.As is described in detail below in section III.2.(b), at the impulseradio receiver a template signal having a waveform designed to match thereceived monocycle pulses is generated. The template signal ispositioned in time according to the known PN code of the transmitter andis then cross correlated with the received impulse radio signal. Thecross correlation output is integrated to recover the impulse radiosignal out of the noise. Once retrieved in this manner, the signal isdemodulated to remove the subcarrier and yield the information signal.

Another embodiment of the impulse radio transmitter according to thepresent invention is shown in FIG. 11. In this embodiment, the positionsof code time modulator 1008 and the subcarrier time modulator 1016 arereversed. As shown in FIG. 11, the information source 1018 outputs theinformation signal 1020 to the subcarrier generator and modulator 1022.In turn, the subcarrier generator and modulator 1022 outputs themodulated subcarrier signal 1024 to the subcarrier time modulator 1016.The subcarrier time modulator 1016 uses the modulated subcarrier signal1024 to time position modulate the periodic timing signal 1004 togenerate a modulated timing signal 1140. Any of the subcarriermodulation techniques described above in connection with FIG. 10 can beused.

The code source 1006 receives the periodic timing signal 1004 forsynchronization and outputs the code signal 1010 to the code timemodulator 1008. The code time modulator 1008 uses the code signal 1010to further time-position modulate the modulated timing signal 1140 tooutput a modulated, coded timing signal 1142. In a similar manner as theembodiment shown in FIG. 10, the modulated, coded timing signal 1142shown in FIG. 11 is provided to the output stage 1028. As describedabove in connection with FIG. 10, the impulse radio transmitter thenoutputs an emitted signal 1012.

The above description of FIG. 11 is exemplary of the many modificationsthat can be made to the impulse radio transmitter to provide thenecessary coding and subcarrier modulation of the signals to betransmitted via the impulse radio transmitter. The above embodimentsdescribed in connection with FIGS. 10 and 11 have been provided by wayof example, not limitation. Similar arrangements of the blocks in FIGS.10 and 11 of the impulse radio transmitter would be apparent to a personskilled in the relevant art based on the above disclosure withoutdeparting from the scope of the invention.

Another transmitter embodiment is shown in FIG. 12. In this embodiment,a summer 1202, or the like, is used to sum the code signal 1010 and aninformation modulated subcarrier signal 1204. The summer 1202 outputs acode modulated subcarrier signal 1206 to a code and timing modulator1208. The code and time modulator 1208 performs the functions of thecode time modulator and the subcarrier time modulator 1016 of FIG. 10.The code and timing modulator 1208 uses the code modulated subcarriersignal 1206 to modulate the periodic timing signal 1004 and thus producethe modulated, coded timing signal 1026. The remaining elements of thereceiver of FIG. 12 operate as discussed in connection with FIG. 10. Anyof the subcarrier modulation techniques described above in connectionwith FIG. 10 can be used.

In a still further alternate embodiment, modulation can be done usingthe information signal 1020 to directly modulate the code signal 1010.This is illustrated in FIG. 13. Summer 1202 is configured to modulate(sum) the code signal 1010 with the information signal 1020 to therebygenerate a modulation signal 1302. A code and timing modulator 1208 usesthe modulation signal 1302 to modulate the periodic timing signal 1004and produce the modulated, coded timing signal 1026. The remainingelements of the receiver in FIG. 13 operate as discussed in connectionwith FIG. 10.

A subcarrier not modulated with information can also be used to modulatethe coded timing signal, or the coded timing signal itself can betransmitted without any modulation. These two latter embodiments can beused to communicate the mere presence of an impulse radio like a beaconor a transponder. Different impulse radio units can be assigneddifferent PN codes and different subcarriers to realize many operationalapplications.

III.2.b. Receiver

An impulse radio receiver 903 for a the single channel subcarrierimpulse radio communication system is now described with reference toFIG. 14.

An impulse radio receiver (hereafter called the receiver) 1400 comprisesa receive antenna 1402 for receiving a propagated impulse radio signal1404. A received signal 1406 is input to a cross correlator 1408 via areceiver transmission line 1410, coupled to the receive antenna 1402.

The receiver 1400 also comprises a decode source 1410 and an adjustabletime base 1414. The decode source 1410 generates a decode control signal1412 corresponding to the PN code used by the associated impulse radiotransmitter (not shown) that transmitted the propagated signal 1404. Theadjustable time base 1414 generates a periodic timing signal 1416 thatcomprises a train of template signal pulses having waveformssubstantially equivalent to each pulse of the received signal 1406. Eachpulse of the received signal 1406 resembles the derivative of a Gaussianmonocycle pulse. FIG. 15 shows a representative plot of a pulse 1502corresponding to the received signal 1406 in connection with thereceiver 1400. The pulse 1502 corresponds to an emitted signal(monocycle pulse) having a waveform like pulse 302 of FIG. 3. When anelectromagnetic monocycle pulse having a waveform like pulse 302 isincident to the receive antenna 1402, the receive antenna has aninherent characteristic that causes the resulting electrical waveform atits output to have the shape of pulse 1502. If the impulse radio antennais inverted, pulse 1502 will be voltage inverted.

FIG. 16 illustrates the cross correlation process. FIG. 16 shows thewaveform of a template signal pulse 1602 and a waveform of a received(impulse radio pulse) signal 1406 at time increments of Δt. A curve 1604is not a continuous waveform, but represents resulting correlationvoltages at each Δt time alignment as the received signal 1406 slides bythe template signal pulse 1602 out of lock. (Note that each Δt of thereceived signal 1406 is voltage inverted when compared to the pulse1502.) The time positioning of the template signal pulse used tocorrelate with the received signal 1406 is established by a decodetiming modulator 1418.

The effect of using the cross correlation function for the modulationtransfer function is to cause the output of the receiver to be anon-linear function of the amplitude of the input. For basebandmodulation, this is undesirable. However, for subcarriers, such as FM,PSK, FSK and Manchester, the harmonics can easily be filtered therebyeliminating any distortion. Such filtering can not remove harmonics whenbaseband modulation is used, because the harmonics stay at baseband, andthus the signal is irrecoverable.

Turning again to FIG. 14, the decode control signal 1412 and periodictiming signal 1416 are received by the decode timing modulator 1418. Thedecode timing modulator 1418 uses the decode control signal 1412 toposition in time the periodic timing signal 1416 to generate a decodesignal 1420. The decode signal 1420 is thus matched in time to the knownPN code of the transmitter so that the received signal 1406 can bedetected in the cross correlator 1408.

The detection process performed by the cross correlator 1408 comprises across correlation operation of the received signal 1406 with the decodesignal 1420. Integration over time of the cross correlation generates abaseband signal 1422. As discussed above in section II.A, integrationover time of the cross correlated signal pulls the impulse radio signalsout of the noise.

In the present embodiment, the baseband signal 1422 is demodulated by asubcarrier demodulator 1424 to remove the subcarrier and yield ademodulated information signal 1426. The demodulated information signal1426 is substantially identical to the information signal of thetransmitter (see 1018 of FIG. 10).

The baseband signal 1422 is also input to a lowpass filter 1428. Acontrol loop 1429 comprising the lowpass filter 1428 is used to generatean error signal 1430 to provide minor phase adjustments to theadjustable time base 1414 to time position the periodic timing signal1416 in relation to the position of the received signal 1406.

The subcarrier embodiments provide less signal compression, and lowersignal distortion by reducing baseband noise for high reliability voice,data and/or imagery communications. The linearity requirements for themodulation using the cross correlator are greatly relaxed by using thesubcarrier technique of the present invention. Modulation transfercharacteristics have to be extremely linear in order to successfullytransfer low distortion speech or music. This is very difficult toachieve in a non-subcarrier baseband impulse system.

Information signals are easily corrupted by noise. Most of the noiseconcentrates at the baseband and then decreases with higher and higherfrequencies, up to the Nyquest frequency. For example, in an impulseradio using a 1.4 megapulse per second rate, the Nyquest frequency wouldbe about 700 kHz. In this example, a subcarrier up to about 700 kHz canbe used to render the impulse radio system substantially impervious tonoise.

In an FM subcarrier embodiment, a phase-locked loop (PLL) frequencydemodulator is used. The characteristics of the phase-locked loopdetermine the bandwidth capture and other basic aspects of the receivedsignal. An optional bandpass filter can be used in series before thephase-locked loop to narrow the spectrum of demodulation performed bythe phase-locked loop.

III.3 Two or More Subcarrier Channels (e.g., voice, digital data andControl Information)

A major advantage of the present subcarrier impulse radio is thatmultiple subcarriers can be packed on the same coded timing signal forsimultaneous transmission. An example of three subcarriers on oneimpulse radio ultrawide-band transmission is illustrated for both analogand digital implementations in FIGS. 17-19.

FIG. 17 shows a representative illustration of an impulse radiotransmitter having three subcarrier generator/modulators (SC GEN/MOD)1702, 1704 and 1706, each having a different subcarrier frequency. Thebasic architecture of the transmitter is based on the embodiment of FIG.10. For example, a main subcarrier generator/modulator 1720 (shown as adashed box) is analogous to subcarrier generator/modulator 1022.However, this example can be modified to operate with any of the abovedisclosed transmitters and their equivalents.

A voice information source (VIS) 1708 is fed to a subcarriergenerator/modulator (abbreviated SC GEN/MOD in FIG. 17) 1702 via a line1722 for modulation of a first subcarrier signal (not shown). The firstsubcarrier signal is internally generated by subcarriergenerator/modulator 1702, or is externally generated and supplied as aninput to subcarrier generator/modulator 1720.

Similarly, a digital data source (DDS) 1710, such as a modem output orfacsimile transmission, is fed to a second subcarriergenerator/modulator (abbreviated SC GEN/MOD in FIG. 17) 1704, via a line(or bus) 1724, for modulation of second subcarrier signals. Finally, adigital control information source (CIS) 1712 is fed to a thirdsubcarrier generator/modulator (abbreviated SC GEN/MOD in FIG. 17) 1706,via a line (or bus) 1726, for modulation of a third subcarrier signal.The second and third subcarriers signals are generated by subcarriergenerator/modulators 1704 and 1706, respectively, or they are externallysupplied as inputs to subcarrier generator/modulator 1720.

The digital CIS 1712 provides control information to an impulse radioreceiver. In a cellular telephone transceiver type system, such digitalcontrol information can comprise routing information, schedulinginformation, ring signals, or the like. Virtually any type of controlsignals, or for that matter, intelligence, can be used to modulate asubcarrier signal.

Three modulated subcarrier signals are output by the three subcarriergenerators/modulators 1702, 1704 and 1706, via lines 1728, 1730 and1732, and are summed at a summer 1714. A resultant signal 1716 is sentto the subcarrier time modulator 1016, where it is used to modulate thecoded timing signal 1014 to generate modulated, coded timing signal1026. The modulated, coded timing signal 1026 output by the subcarriertime modulator 1016 is fed to the output stage 1028 and transmitted asan emitted signal 1012 as described above.

Two representative plural subcarrier channel impulse radio receivers areshown in FIGS. 18 and 19. Each receiver has components for demodulatingthe three subcarrier channels transmitted by the transmitter of FIG. 17,for example. The basic architecture of the receivers in FIGS. 18 and 19is based on the embodiment of FIG. 14, or its equivalents.

FIG. 18 is a representative analog embodiment showing the crosscorrelator 1408 followed by plural analog FM demodulation branches. Thecross correlated baseband signal 1422 is generated from the receivedsignal 1406, as discussed in connection with FIG. 14 (using the elementsof the control loop which are not illustrated in FIGS. 18 and 19). Eachbranch demodulates one subcarrier using a bandpass filter 1802 (e.g., anL-C or switched capacitor filter) and a phase-locked loop block 1804.Thus, three separate, simultaneously transmitted information signals arerecovered and made available at OUTPUTs 1-3.

In a digital embodiment shown at FIG. 19, the cross correlated basebandsignal 1422 is converted into a digital signal using ananalog-to-digital converter (A/DC) 1902. Using a digital signalprocessor (DSP) 1904, such as a model no. TMS320C40 DSP package(manufactured by Texas Instruments, Dallas, Tex.), or the like, andknown digital signal processor algorithms using Fourier transforms orthe like, the three separate subcarriers encoded in signal 1903 aredigitally demodulated. The digitally demodulated information can beconverted back to analog using digital-to-analog converter (D/AC) 1906.The voice signal is converted back to its analog counterpart usingdigital-to-analog converter 1906 and made available at OUTPUT 1. Thedigital data signal is output or otherwise made available directly fromthe digital signal processor at OUTPUT 2. Finally, the control signal isoutput or otherwise made available at OUTPUT 3 directly from the digitalsignal processor or after digital-to-analog conversion bydigital-to-analog converter 1906. The addition of plural subcarriersdoes not affect the wide band characteristics of the impulse radiosignals.

IV. The Time Modulator

This section is directed to the time modulator that is used for codetime delaying, subcarrier time delaying and a combination of both. Theoperation and structure of several embodiments for using the timemodulator for subcarrier impulse radio communications are described.

In accordance with various embodiments of the present invention, theimpulse radio transmitter includes code time modulators (e.g., 1008) andsubcarrier time modulators (e.g., 1016), as well as code and timingmodulators (e.g., 1208). Each of these modulators functions to timedelay a signal (e.g., the periodic timing signal 1004) according toinformation conveyed by a trigger signal (e.g., code signal 1010 ormodulated subcarrier signal 1024.) Thus, each modulator (e.g., 1008,1016 or 1208) is therefore considered a delay generator. Delaygenerators having numeric input signals are called binary-to-time delaygenerators.

Binary-to-time delay generators can be implemented using currentlyavailable commercial ICs. A preferred delay generator having a numericinput is a model MC100E196 ECL (emitter coupled logic) devicemanufactured by Motorola, of Schaumburg, Ill. However, in connectionwith the impulse radio signals according to the present invention, suchconventional binary-to-time delay generators do not provide accuratetime delays to permit accurate recovery of impulse radio signals at theimpulse radio receiver. In other words, time delays on the order of 157ps (picoseconds), which is a typical pulse duration of a monocyclepulse, cannot accurately be produced using conventional binary-to-timedelay generators.

V. Linearization

This section is directed to linearization of the time modulator for boththe impulse radio transmitter and receiver. Linearization of the timemodulator permits the impulse radio transmitter and receiver to generatetime delays having the necessary accuracy for impulse radiocommunications.

In order to solve the time delay problem described above in section IV.,the inventors have performed statistical analysis of the specifications(e.g., performance curves) provided by binary-to-time delaymanufacturers. Based on this work, the inventors discovered that thenon-linear operational characteristics of conventional binary-to-timedelay generators can be compensated for if the non-linear operationalcharacteristics of the device are known. Thus, according to a furtheraspect of the present invention, the impulse radio transmitter comprisesa linearization look-up read only memory (ROM) (not illustrated), inconjunction with conventional binary-to-time delay generators tocompensate for any non-linearity. This permits the impulse radiotransmitter to generate time delays having accuracy well below the 157ps requirement.

FIG. 20 is a plot showing delay time (in picoseconds) versus a binary(i.e., numeric) input value for a conventional binary-to-time delaygenerator. A curve 2002 shows an example of the actual time delay outputcharacteristics of a conventional binary-to-time delay generator. Thedesired output of a binary-to-time delay generator for use with thepresent invention is shown at a curve 2004.

For a binary input value of 18, for example, a point 2010 on curve 2002represents the actual output of a conventional binary-to-time delaygenerator. A binary input value of 10 would typically be input toproduce a 157 ps time delay at the output of the conventionalbinary-to-time delay generator. However, given the numeric input valueof 10, a conventional binary-to-time delay generator may produce anactual output value of only approximately 15 ps, rather than the desired157 ps, as shown at a point 2006. Thus, in order to generate a 157 psdelay in this example, a numeric input value of 18 would need to beinput to produce the desired delay of 157 ps, as shown at a point 2010on the curve 2002.

Although it is generally desirable to linearize the dither generators onthe transmitter and receiver, it is actually necessary only to have thesame dither versus numeric input mapping linearity, not necessarily astraight line.

According to the present invention, linearization data, of the typeshown in FIG. 20, is used to map the actual response of a conventionalbinary-to-time delay generator to a desired time delay. Thislinearization data, or map, is stored in a linearization read onlymemory (ROM).

In order to transmit 1's and 0's, pulses are time-modulated eitherforward or backward in time. In other words, impulse radio signals, thatare intended to produce a logical value of 1 when received by theimpulse radio receiver, are time positioned slightly forward by theimpulse radio transmitter. Impulse radio signals that are intended to bereceived as logical 0's are time shifted slightly back by the impulseradio transmitter.

The cross correlator 1408 in the impulse radio receiver converts thattime position into more positive or more negative voltage increments. Abandpass data filter is used to maximize the signal-to-noise ratio ofthe data stream. The bandwidth of this bandpass data filter is set toapproximately one-half the transmission baud rate, as would be apparentto a person skilled in the relevant art. A comparator then turns thosevoltages into logical equivalents of 1's and 0's. It is necessary tosupply a pulse for both 1's and 0's because, in the absence of a pulse,noise at the threshold of the comparator would produce a random output.The larger the separation (i.e., voltage difference) between thepositive and negative information samples, the better the signal tonoise ratio and the lower the bit error rate.

Because the 1's and 0's cause the signal to be time shifted, thelinearization ROM must store separate linearization information forimpulse radio signals for the logic 1 and separate linearization datafor impulse radio signals for the logic 0. For a predeterminedinformation (data) transmission rate, impulse radio transmission logic1's and logic 0's must be shifted ahead and back, respectively, in timeby a finite amount so that the cross correlator in the impulse radioreceiver can properly distinguish logic 1's from logic 0's in the datastream.

For a chosen center frequency of the monocycle pulse of 1.3 GHz, forexample, the desired shifting forward for logic 1's and shiftingbackward for logic 0's is a shift value of 157 ps; if the centerfrequency doubles, the time shift is halved. Thus, linearization ROMmust store one (8 bit) digital value representing a linearized numericvalue, such that when output from the linearization ROM to the code timemodulator 1408, the proper 157 ps time shift can be realized. In apreferred embodiment, the linearization ROM will store one 8 bitnumerical value for a forward shift of 157 ps and a second 8 bitnumerical value for a backward shift of 157 ps. In order to achieveforward and backward shifts of some other time shift in addition to thatof 157 ps, the linearization ROM must store further 8 bit numericalvalues for forward and backward time shifts. Note that if thetransmitter used a modulator employing a zero time shift (nominal) andtwo times 157 ps for the modulation values (corresponding to digitalzero and one, respectively), that this would look the same to ademodulating receiver.

V.1. Transmitter

FIG. 21 is a high-level block diagram showing the above linearizationscheme according to the present invention. However, in contrast to thecoded timing signal 1014 generated by the code time modulator 1008 ofFIG. 10, for example, a direct digital coded timing signal 2102 isproduced by the code time modulator 1008 as illustrated in the blockdiagram of FIG. 21.

In this embodiment, the time base 1002 outputs the periodic timingsignal 1004 to the code source 1006. The periodic timing signal 1004 isalso provided to the code time modulator 1008, which in this embodimentis a binary-to-time delay generator.

In this embodiment the code source 106 comprises an address counter 2104and two read-only memories (ROMs) 2106 and 2110.

The periodic timing signal 1004 increments the address counter 2104 thatoutputs a multi-bit address 2105. In this example embodiment, theaddress counter 2104 outputs a 15-bit-wide address 2105 for each pulseof the periodic timing signal 1004.

The address 2105 provided by address counter 2104 is used to access a PNcode ROM 2106. The ROM 2106 stores PN (pseudo-random noise) code of apredetermined modulo. (Alternatively, other memory devices such as anEEPROM, RAM, shift registers, or the like can be used.) Each address2105 output from the address counter 2104 accesses a storage location inthe ROM 2106, which in response thereto, outputs a PN code 2108(preferably a 15-bit PN code). (As described above, the PN codes areused to time-position modulate pulses (e.g., periodic time signal pulseor digital data signal pulses) ahead or back in time for channelizationand spreading of the monocycle pulses of the impulse radio signal.)

Linearization data is stored at addressable locations in a linearizationROM 2110. The linearization data is accessed by application of anaddress (e.g. a 16-bit address) to address inputs of the linearizationROM 2110. According to a preferred embodiment of the present invention,the 16-bit address is formed by the concatenation of the 15-bit PN code2108 output by ROM 2106 and a 1-bit digital data source (shown by dashedline 2107, which is analogous to 1024 of FIG. 10) provided byinformation source 1018, for example.

Alternatively, the digital data provided by information source 1018 canbe used to modulate a subcarrier using the subcarrier/generator 1022, asdescribed herein. In this case, the subcarrier/generator 1022 wouldprovide the 1-bit digital data signal (see solid line 2109) to thelinearization ROM 2110.

In response to synchronized receipt of a complete input address (16 bitsin this example), the linearization ROM 2110 outputs a linearized,modulated timing signal 2112 (which is analogous to 1206 of FIG. 12 and1302 of FIG. 13). The linearized, modulated timing signal 2112 ispreferably 8-bits wide and is provided to the code time modulator (i.e.,binary-to-time delay generator) 1008. The code time modulator 1008 usesthe modulated timing signal 2112 to time delay the periodic timingsignal 1004 and thus output the direct digital coded timing signal 2102.

The linearization ROM 2110 stores linearization data in order toproperly linearize time delays provided by the PN code ROM 2106. Each 15bit pseudo-random code 2108 provided to the linearization ROM 2110represents a dither time delay used to time modulate the digital databit 2107 that is simultaneously provided to the linearization ROM 2110.In this embodiment, 2¹⁵ (23,768) different time delays can be used totime modulate the forward time shift of logic 1 or the backward timeshift of logic 0. The modulation of the time delay composed by the PNnoise code prior to cross correlation in the impulse radio receiverpermits recovery of the data. The preferred embodiment of the impulseradio receiver describing this operation is discussed below.

FIG. 22 is a functional diagram illustrating linearization ROM 2110.Locations 2202 and 2204 of the FIG. 22 represent storage locationswithin the linearization ROM 2110 addressed by high-order addresses andlow order addresses, respectively. In this example, each storagelocation can store 8 bits of data. Thus, in this example, the datastored within the linearization ROM 2110 is separated in two groups: thedata in locations 2202 and the data in locations 2204. The first groupof data (locations 2202) represents linearization data used when digitaldata source 2107 is a logic 1, for example, and the linearization datastored in the second group (locations 2204) represent linearization dataused when digital data source 2107 is a logic 0. Thus, the logic valueof the digital data source 2107 which forms the most significant bit ofthe ROM address dictates whether linearization data will be output fromblocks 2202 or from blocks 2204.

The 15 bits of PN code 2108 applied to the 15 least significant addressinputs of the linearization ROM 2110 are used to select which specificROM location within either selected set of locations 2202 or 04 will beoutput by linearization ROM 2110.

In a further embodiment of the present invention, the PN codes can bemathematically combined with the linearization data and the resultantnumeric information can be stored directly in a single ROM, or the like.This further embodiment avoids the need for two ROMs. The addresscounter 2104 would simply directly input addresses to a single PNcode/linearization ROM. (In spread spectrum theory, each element of a PNcode is called a “chip.” Thus, a PN code having a length of modulo Ncomprises a total of N chips.) Rather than the first ROM outputting adesired delay value for each code chip and then linearize each delayvalue, a single ROM can be used to store a linearized version of thedesired delay for each code chip.

A still further embodiment of the impulse radio transmitter is shown inthe block diagram, FIG. 23. In FIG. 23 a combined PN code andlinearization E-PROM 2302 is used to generate an 8-bit coded informationsignal 2304, which represents a time delay to be generated by the codetime modulator 1008. Use of the PN code can be switched on and off usinga code switch 2306. The code may be eliminated for various reasons, suchas a separate operational mode that permits accelerated signalacquisition and lock at the impulse radio receiver. The code switch 2306can be controlled by a simple switch, separate control logic, amicroprocessor, or the like. With the code switched on, as shown in FIG.23, the time base 1002 is used to clock the address generator 2104, asdescribed above in connection with FIG. 21. However, in FIG. 23, thetime base is shown as being implemented with a VCO 2308 and aprogrammable divider 2310. The functions performed by the VCO 2308 andthe programmable divider 2310 would be apparent to a person skilled inthe relevant art.

In accordance with the embodiment illustrated in FIG. 23, a counterstart page block 2312, a counter stop page block 2314 and a counterlimit comparator block 2316 are included. The counter start page block2312 provides an address (preferably 15 bits) to the address generator2104 to indicate a starting address. The counter stop page block 2314provides an address (also preferably 15 bits) to the counter limitcomparator block 2316 to indicate a stop address. The counter limitcomparators of block 2316 comprise logic to compare the address,generated by the address generator 2104, to the stop page addressprovided by counter stop page 2314. The counter limit comparators block2316 generates a load signal 2317 and forwards the load signal 2317 tothe address generator 2104 when a comparison of these addresses yieldsan equality. In response to receipt of the load signal 2317, the addressgenerator 2104 is reset and begins counting again at the 15-bit addressspecified by the counter start page 2312. The process of counting upfrom the start page address to the stop page address is repeatedcontinuously. The repeating of these addresses permits the PN code andlinearization E-PROM 2306 to modulate the digital data with a PN codemodulo of a length determined by the difference between the counterstart page and the counter stop page addresses.

As noted above, combined PN code and linearization E-PROM 2302 is usedto generate an 8-bit coded information signal 2304, which represents atime delay to be generated by the code time modulator 1008. Code timemodulator 1008 time position modulates the coded information signal 2304using the periodic timing signal 1004. The code time modulator 1008outputs the direct digital coded timing signal 2102, as described abovein connection with FIG. 21.

The embodiment illustrated in FIG. 23 also includes an FM subcarriermodulator 2318. The FM subcarrier modulator 2318 generates a sinusoidalsignal 2320. The sinusoidal signal 2320 is summed with baseband audiosignal 342 provided by baseband audio source 2344 at a summer 2322. Notethat baseband audio source is an example of information source 1018.

The summer 2322 outputs a modulator signal 2324 used by the subcarriertime modulator 1016 in a manner similar to that described above inconnection with FIG. 10. When decoded by the impulse radio receiver, therecovered sinusoidal signal 2320 can be used as a control signal by theimpulse radio receiver. Thus, the embodiment of the impulse radiotransmitter illustrated by FIG. 23 transmits three separate informationconveying signals in a single impulse radio transmission. These threeinformation conveying signals comprise the digital data 2107, thesinusoidal signal 2320 and the baseband audio signal 2342.

Alternatively, block 2344 in FIG. 23 can be replaced by a subcarriergenerator and modulator 1022, as described above in connection with FIG.10, or blocks 1018 and 2318 could each be replaced with one of thesubcarrier generators/modulators 1702, 1704, 1706, as described above inconnection with FIG. 17.

According to still a further embodiment, the direct digital coded timingsignal 2102 can be directly input to the output stage 1028. In thisembodiment, Manchester coding is the only form of subcarrier modulationperformed. Other configurations will be apparent to a person skilled inthe relevant art after reading this disclosure.

V.2. Receiver

A further embodiment of the impulse radio receiver is illustrated inFIG. 24. This embodiment of the impulse radio receiver is similar inmany respects to the receiver described above in connection with FIG.14. The receiver illustrated in FIG. 24 comprises a cross correlator1408, a subcarrier demodulator 1424, a low pass filter 1428, anadjustable time base 1414, a decode timing modulator/decode source 2402,a pseudo Manchester decoder 2404 and a microprocessor 2406.

According to this embodiment, a propagation signal (1404) is received bythe impulse radio receiver antenna 1402, which passes the receivedsignal 1406 to an RF amplifier 2408. The RF amplifier 2408 amplifies andpasses the received signal to the cross correlator 1408.

The cross correlator 1408 can include a multiplier 2410, a triggeredwaveform generator 2412, an amplifier 2414, an integrator 2416, a sampleand hold unit 2418, and a delay unit 2420. The multiplier 2410 is adouble balanced mixer adapted to operate in the linear mode. Themultiplier 2410 linearly multiplies the received signal with a templatesignal 2422 generated by the triggered waveform generator 2412. Aproduct signal 2415 of the multiplier 2410 is buffered by amplifier 2414and then integrated over time by integrator 2416. The integrator isessentially a low-pass filter of first order, which is adapted torespond on a time scale similar to the width of the monocycle (i.e., 157ps). Integrator 2416 outputs a signal 2417 to the sample and hold unit2418 that holds the peak value of signal 2417.

The delay unit 2420 is for proper triggering of the sample and hold unit2418. The delay unit 2420 allows for delay caused by the multiplier2410, and the amplifier 2414, and for integrator settling time. In oneembodiment, the delay unit 2420 delays triggering approximately 10-25 nsafter the peak value produced by the integrator 2416. As a result,sampling occurs before the integrated value degrades.

According to this embodiment of the impulse radio receiver, the decodesignal 1420 is generated in a manner similar to generation of the directdigital coded time signal 2102, discussed above in connection with FIG.21. The main difference for block 2402 in the impulse radio receiver,versus the impulse radio transmitter, is that a data source is not usedto access the PN code/linearization ROM.

Decode timing modulator/decode source 2402 comprises a binary-to-timedelay generator 2424, a PN code and linearization ROM 2426, and anaddress counter and limit logic block 2428. Start address and stopaddress signals are provided to the address counter and limit logicblock 2428 from the microprocessor 2406 via lines 2430 and 2432,respectively. Addresses are output from the address counter and limitlogic block 2428 via a bus 2434. The address counter and limit logicblock 2428 provides addresses to access the PN code and linearizationROM 2426 when triggered by the periodic timing signal 1416 provided bythe adjustable time base 1414. A PN code (that corresponds to a known PNcode used by an impulse radio transmitter) is output by the PN code andlinearization ROM 2426 via a bus 2436 and is provided to thebinary-to-time delay generator 2424. The binary-to-time delay generator2424 time modulates the periodic timing signal 1416 (received fromadjustable time base 1414) to generate the decode signal 1420.

In this example, the adjustable time base 1414 comprises a programmabledivider 2438 and a voltage controlled oscillator (VCO) 2440, which areused to output the periodic timing signal 1416. A voltage control signalis provided to the VCO 2440 from the microprocessor 2406 via a line 2442to adjust the VCO output, as will be apparent to a person skilled in therelevant art.

In this example, the subcarrier demodulator 1424, comprises a bandpassfilter 2444, a phase-locked loop 2446, and a low-pass filter 2448. Thefunction performed by the phase-locked loop 2446 is equivalent to thatperformed by similar phase-locked loops (2004) in FIG. 18. Similarly,the bandpass filter 2444 performs a similar function as the band filters1802 in FIG. 18. In this case, bandpass filter 2444 outputs a filteredsignal 2445 to the phase-locked loop 2446. The phase-locked loop 2446outputs an in-phase estimate signal 2447 via a further low pass filter2449 to the microprocessor 2406. The in-phase estimate signal 2447provides the microprocessor 2406 with an estimate of the amplitude ofthe subcarrier so that the microprocessor 2406 can assess the quality ofsignal lock. A demodulated output signal 2450 of the phase-locked loop2446 is filtered by low-pass filter 2448, which in turn outputsdemodulated information signal 1426.

The overall functionality and operation of the subcarrier demodulator1424 in FIG. 24 is substantially the same as that described above inconnection with FIG. 14. The control loop 1429 has the samefunctionality as described above in connection with FIG. 14.

Additional subcarrier modulation is achieved according to another aspectof the invention using pseudo Manchester coding of digital data. It isreferred to as “pseudo” because conventional Manchester coding performsdigital decoding. According to the present invention, however, decodingof Manchester encoded signals is performed in the analog domain. Thepseudo-Manchester encoding shifts digital information from the basebandto a frequency equivalent to an integral submultiple of the adjustabletime base, or integer multiples of the time base. This achieves acoherent shift of digital data for proper recovery in the impulse radioreceiver.

In this embodiment, the pseudo Manchester decoder 2404 comprises abandpass filter 2450 and an analog Manchester decoder 2452. The bandpassfilter 2450 receives the baseband signal 1422 from cross correlator1408. A filtered baseband signal 2454 is provided to analog Manchesterdecoder 2452. The decoding performed by the analog Manchester decoder2452 is best described after explanation of the actual encodingperformed at the transmitter.

Additionally, various signals numbered in FIG. 24 are illustrated astime (t) versus voltage plots in FIGS. 25A-25H. Additional FIGS. 25I-25Lare frequency versus amplitude (P or log P) plots that correspond toFIGS. 25E-25H.

VI. Pseudo Manchester Modulation

This section is directed to pseudo Manchester coding for modulation ofdigital data using impulse radio communications.

Using the direct digital modulation approach, described above inconnection with FIG. 24, a problem may arise when the data sourcegenerates a long string of logic “1's” or logic “0's”. Because the datais recovered using the phase-locked loop, the low-frequency energy insuch a string of “1's” or “0's” appears in the low-pass filter 1428 thusintroducing a phase error in the error loop 1429. A method of separatingthe modulation frequency components from those expected in the errorloop 1429 is necessary.

Accordingly, the inventors have developed further subcarrier embodiment.This further subcarrier embodiment comprises a modulation scheme inwhich data is synchronously exclusive-ORed (XORed) with a square wavewhose frequency is at least two times the frequency of the data signal(a 2×clock) in the manner of Manchester coding. Use of Manchester codingin a impulse radio system is a subcarrier technique because the data ismodulated to a higher frequency using the 2×clock.

The impulse radio receiver removes this modulation in an analog fashionrather than digitally, as is done with true Manchester decoding. Thevoltage from the sample-and-hold (2418) is modulated by a synchronous2×clock, and sequentially processed by a low-pass filter followed by acomparator (not shown). The simplest embodiment is a low-pass filter setto cut off at a frequency above approximately half the bit rate;however, more complex filtering would be needed for use with othertranslation methods. Thus, the pseudo Manchester modulation technique,according to this aspect of the present invention, convertsnon-return-to-zero (NRZ) digital signals to return-to-zero (RZ) signalsto avoid errors in the phase-locked loop of the impulse radio receiver.The return-to-zero encoder can be a pseudo Manchester direct digitalencoder, a frequency shift keying encoder, an n-ary phase modulationencoder (e.g., quadrature phase shift keying (QPSK)) or a phaseamplitude modulation encoder, or other frequency translation means aswould be apparent to a person skilled in the relevant art.

The pseudo Manchester coding scheme according to the present inventionuses a standard implementation of Manchester coding in the impulse radiotransmitter. The digital data stream is Manchester coded before it isused to address the PN-code and linearization E-PROM 2302, for example(see FIG. 23). The circuitry to implement Manchester coding of thedigital data stream would be apparent to a person skilled in therelevant art.

FIGS. 26 and 27 show exemplary waveforms for pseudo Manchester encodingand decoding, respectively, according to the present invention. In FIG.26, a sample digital data stream of logic 1's and 0's is shown generallyat a waveform 2602. In the impulse radio transmitter, the data is XORedwith a square wave whose frequency is at least two times the frequencyof the data signal (a 2×clock), as shown at a waveform 2604. Thewaveform 2604 must be synchronous and have transitions aligned with adata bit edge. The XORed result of waveforms 2602 and 2604 is showngenerally at a waveform 2606. This process ensures a zero-to-one orone-to-zero transition at the middle of each bit period, whicheliminates problems associated with a long run of 1's or 0's.

In connection with the pseudo Manchester coding embodiment, the impulseradio receiver performs pseudo Manchester decoding to recover thedigital data signal. A set of waveforms illustrating the functionsperformed to recover the data are shown in a FIG. 27. Once a receivedimpulse radio signal is cross correlated in the impulse radio receiver,it is passed through the bandpass filter 2450. The output 2454 ofbandpass filter 2450 resembles an exemplary waveform shown generally at2702.

The filtered baseband signal 2454 is fed to a first input of an analogmultiplier (not shown). The second input of the analog multiplierreceives a synchronous, 2×clock signal (2704). The analog multiplierreverses the process performed by the impulse radio transmitter. Aproduct (output) signal of the analog multiplier is shown generally at awaveform 2706. The product signal is low-pass filtered and compared witha predetermined comparison level, as shown generally by the waveform at2708, thus yielding a compare data signal 2710. The compare data signal2710 is held at the peak of the impulse/response point of the filter(using a sample and hold unit), via the rising edge of a data strobesignal (shown generally at waveform 2712), to produce recovered data2465, shown generally at a waveform 2714. The analog recovery technique,according to the present invention, takes advantage of the coherentlink, i.e., synchronous recovery, to reduce the noise as much astheoretically possible with filtering.

VII. Lock Acquisition Scheme

This section is directed to a lock acquisition scheme for the impulseradio receiver to acquire and maintain lock of impulse radio signals.

As with all communications receivers, the impulse radio receiver mustfirst acquire and maintain a “lock” on the signal, before data can berecovered. FIG. 28 is a high level block diagram of operations performedby the impulse radio receiver to acquire lock.

Once the transmitter and receiver are turned on, as shown at steps 2802and 2804, respectively, the microprocessor 2406 applies bias to the VCO2440 (as shown at a step 2806) to cause the lock loop 1429 to drift at aprogrammed rate faster (or slower) than the remote transmitter'stransmit period, as shown at a step 2808. A few parts-per million is atypical offset.

Next, the microprocessor 2406 digitizes the voltage from the crosscorrelator 1408 (received via the filter 1428) looking for a non-zeroaverage voltage, which indicates the template signal is in nearalignment with the received signal, as shown at a step 2808. Themicroprocessor then reduces the difference in rates (receive VCO vs. thetransmitter VCO) to begin scanning the time around the perceived timethat the energy was detected, as shown at a step 2810.

Alternatively, the digitizing performed by the microprocessor 2406 couldbe done using separate A/D converter hardware. Similarly, filteringcould be done by the microprocessor, discrete components or activefilters, as would be apparent to a person skilled in the art.

When the time corresponding to the maximum correlation energy isdetected, the microprocessor switches to a tracking algorithm in whichthe correlator's average voltage is kept to zero, as shown at a step2812. This tracking is analogous to the quadrature lock algorithm usedin conventional CW phase locked loop designs, as would be apparent to aperson skilled in the art. Thus, once the tracking algorithm is engaged,subcarrier demodulation of data by the pseudo Manchester decoder 2404and the subcarrier demodulator 1424 can begin.

VIII. Performance in the Real World

This section describes the performance of the impulse radiocommunications system in the real world with reference to data collectedby the inventors based prototype testing of impulse radio systems.

One impulse radio prototype, built by the inventors, has an averageradiated power of 450 microwatts (μW). The center frequency is 675 MHzand smoothed by a pseudo-random code with 256 positions. FIG. 29 showsthe signal measured at 3 meters (see plot 2902) as well as ambientsignals (see plot 2904). Measurements for this figure were not adjustedto compensate for antenna performance and a 1.3 GHz/2 mpps prototype wasused with an average output power of 33 μW. A power spike 2906 justbelow 900 MHz is from two cellular base stations, one about 400 metersdistant and another about 1.6 kilometers distant. Spikes 2908 between360 MHz and 720 MHz are predominantly UHF televisions stations. The 720MHz spike is a 2.2 megawatt EIRP channel 54 station, Huntsville, Ala.,approximately 7 miles distant. (The “bumpiness” of the impulse spectralmeasurements reflects the impact of frequency domain multipath. Movingthe receive antenna causes the location of nulls and peaks to move. Thisdoes not impact the performance of the impulse system.)

Impulse radio performance has been measured for a 1.3 GHz/2 mppsprototype (with an average output power of 33 μW) over two paths:

-   -   1) With a −9.6 dBi transmit antenna buried in a highly        conductive medium having a total loss of 36 dB over a 6 cm path,        the inventors used the impulse radio to transmitted a 125 kbps        pseudo-random bit stream an additional 4 meters through air to a        10 dB_(i) receive antenna. The bit error rate was better than        0.5×10⁻⁵.    -   2) With the same experimental set-up and the same location, the        bit rate was lowered to 7.8 kbps and range was increased to 10        meters. The bit error rate was better than 10⁻⁶.

One can project the performance of the 1.3 GHz/2 mpps simplex link infree space using standard propagation modeling assumptions. FIG. 30shows a curve 3002 that illustrates the projected trade-off between freespace range and bit rate, assuming a 100 μW average power (−10 dB_(m)),a 10 dB_(i) receive antenna (approximately a 90° beam), a 2 dB_(i)transmit antenna (omni-directional dipole-like pattern), an SNR of 19.5dB (approximately a 10⁻⁶ BER), and a margin of 6 dB.

Turning to FIG. 31, this figure shows that it is easy to resolvemultipath impulse signals in the time domain. Measurements illustratedat plot 3102 were made in a laboratory in a single story office complex.The laboratory contained many feet of steel shelving, test equipment,and metal filing cabinets. One adjacent office space is occupied by ametal fabricating company. The other is occupied by a personal computersales offices along with that company's warehouse (using steelshelving).

The first arriving pulse (between 3 ns and 6 ns) is of lower amplitudebecause it traveled through more walls than some later arriving pulses.

IX. Conclusion

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample, and not limitation. Thus the breadth and scope of the presentinvention should not be limited by any of the above-described exemplaryembodiments, but should be defined only in accordance with the followingclaims and their equivalents.

1. A transmitter, said transmitter comprising: (a) an output stage thatgenerates a monocycle pulse; (b) a filter that spectrally modifies themonocycle pulse to create a spectrally modified ultra wideband signalhaving more zero crossings than the monocycle pulse in the time domain;and (c) an antenna coupled to said filter that radiates said spectrallymodified ultra wideband signal.
 2. The transmitter of claim 1, whereinsaid filter is a bandpass filter.
 3. The transmitter of claim 1, whereinsaid output stage generates said monocycle pulse based upon a triggersignal.
 4. The transmitter of claim 3, wherein said trigger signal isbased on at least one of an information signal, a code signal, and asubcarrier signal.
 5. A method of transmitting, comprising: (a)generating a monocycle pulse; (b) spectrally modifying the monocyclepulse to create a spectrally modified ultra wideband signal having morezero crossings than the monocycle pulse in the time domain; and (c)radiating the spectrally modified ultra wideband signal.
 6. The methodof claim 5, wherein a filter is used to spectrally modify the monocyclepulse.
 7. The method of claim 6, wherein said filter is a bandpassfilter.
 8. The method of claim 5, wherein said generating the monocyclepulse is based on a trigger signal.
 9. The method of claim 8, whereinsaid trigger signal is a based on at least one of an information signal,a code signal, and a subcarrier signal.
 10. A method of transmitting,comprising: (a) generating a monocycle pulse; (b) filtering themonocycle pulse to create a filtered ultra wideband signal having morezero crossings than the monocycle pulse in the time domain; and (c)radiating the filtered ultra wideband signal.
 11. The method of claim10, wherein said filtering is by a bandpass filter.
 12. The method ofclaim 10, wherein said generating the monocycle pulse is based on atrigger signal.